Method using digital phase-locked loop circuit including a phase delay quantizer

ABSTRACT

A phase locked loop circuit and method for use, in accordance with an embodiment, implements a digital phase delay quantizer to replace the analog charge-pump and phase frequency detector in an analog PLL circuit. Therefore, the built-in loop filter can be a compact-sized, high order, high bandwidth, and high attenuation digital filter as well. The digital PLL circuit takes advantage of the deep sub-micron process technology which features high speed, high resolution, compact size, and low power.

CROSS-REFERENCE TO RELATED APPLICATIONS

Under 35 U.S.C. 120, this application is a Divisional Application andclaims priority to U.S. application Ser. No. 12/465,547, filed on May13, 2009, which is incorporated herein by reference in its entirety.

FIELD OF THE INVENTION

The present invention relates generally to phase locked loop circuits,and more particularly to a method for providing phase delays for suchcircuits.

BACKGROUND OF THE INVENTION

Phase locked loop (PLL) circuits are used in a variety of applications.FIG. 1 is a diagram of a conventional analog PLL circuit 100. The VCOfrequency f_(VCO) is divided by a frequency divider 112 to get thedivided VCO frequency f_(VCO)/N. A flip-flop based phase frequencydetector 104 compares a reference clock “f_(ref)”, obtains the dividedVCO frequency “f_(VCO)/N” and sends charge-up and charge-down signals toadjust the analog charge-pump 106. The charge-pump 106 adjusts thecontrol voltage up or down based on the phase frequency detector 104comparison results. The charge-pump 106 output voltage is low passfiltered by analog filter 108 and sent to the VCO 110 so as to tune theVCO frequency.

Conventional Analog PLL Circuit function

In the conventional analog PLL circuit 100 shown in FIG. 1, theflip-flop based phase frequency detector 104 compares the frequencies ofreference clock “f_(ref)” and the divided VCO frequency “f_(VCO)/N” toadjust the charge-pump circuit 106 so as to tune the VCO frequency.

FIG. 2 illustrates a slower VCO adjusted by the conventional analog PLLcircuit 100. If the VCO frequency is slower than expected, the dividedVCO frequency f_(VCO)/N arrive later than the reference clock “f_(ref)”.The phase frequency detector 104 sends a longer charge-up time signaland another shorter charge-down time signal to the analog charge-pump106. Consequently, a positive net charge is delivered to the analogfilter 108 from the charge-pump 106, which means the VCO 110 inputcontrol voltage goes up. Finally, a higher control voltage speeds up theVCO 110.

However, as device sizes become smaller such as in deep submicrontechnology, there are problems with this conventional analog design.Namely: (1) a relative large loop analog filter and (2) a low powersupply headroom. The ways for overcoming these issues usually causeadditional problems, which are described below.

1. Large loop filter size in analog PLL circuit solutions.

a) Provide a large built-in passive loop filter. The problem with thistype of filter is that the filter dominates the silicon size and becomesproblematic when utilized in the deep sub-micron process technology dueto size conventions of the

PLL circuit 100.

b) Provide a built-in active loop filter. The problem with this type offilter is that the filter consumes a large amount of power and alsocreates a large amount of noise.

c) Provide a large passive loop filter outside the chip. The problemwith this type of filter is that the filter provides a low integrationlevel and also adds a package interface noise interference component.

2. Low power supply headroom.

a) When using the above-identified topology in smaller processtechnologies, the tunable range, noise, and linearity performances aresacrificed due to the low voltage supply headroom of the analogcharge-pump.

b) Another solution is to use auxiliary circuits to correct for thetuning range, noise, and linearity issues of the charge-pump. Theproblem with using auxiliary circuits is that the circuits increase thesize, power, and complexity of the design; and also the auxiliarycircuits may become the source of additional noise and non-linearity.

To address the above-identified issues, digital PLL circuits have beenimplemented. FIG. 3 illustrates one embodiment of a conventional digitalPLL circuit 200 that attempts to address some of the above-identifiedissues related to analog PLL circuits.

The conventional digital PLL circuit 200 of FIG. 3 utilizes aTime-to-Digital converter (TDC) 205 to replace the analog charge-pump sothat the other blocks can be implemented in digital. The digital PLLcircuit 200 does not require a frequency divider. The DCO high frequencyoutput is sent directly to the TDC to form a feedback loop.

There are several problems with the digital PLL circuit 100 which aredescribed below.

1. The coarse resolution (one delay time of the inverter) of the TDClimits the phase noise and jitter performances.

2. The TDC limited length limits the PLL locking range.

3. The over-sampling design consumes huge power and limits the operatingfrequency of the DCO 210 (PLL output frequency).

Accordingly, what is needed is a system and method for addressing theabove-identified issues.

The system and method should avoid using analog circuits that dominatethe silicon size. In the new deep sub-micron process technologies, thesizes of analog circuits do not shrink as digital counterparts do. Thenon-shrunk analog circuit size implemented with expensive new technologywill increase the cost of the chips. For example, the analog charge-pumpand loop filter dominate the size of conventional PLL circuit.

The low power supply voltage of the deep sub-micron process technologysuppresses the headroom of circuits. The low headroom issue degrades theperformance of analog circuits. The interface between phase frequencydetector and high voltage analog charge-pump in an analog PLL circuithas a voltage level shift issue as well, which degrades the linearityand noise performance. Accordingly, interfaces between analog devicesand digital devices that degrade the performance should be minimized.

Accordingly, a system and method is needed to address theabove-identified issues. The present invention addresses such a need.

SUMMARY

A method quantizing a phase delay is disclosed. In one aspect, a methodincludes providing a reference clock signal, providing filtered digitalcodes by a digital filter, and receiving the filtered digital codes andproviding an output signal. The method also includes receiving theoutput signal and providing a frequency divided signal, quantizing aphase delay between the reference clock signal and the frequency dividedsignal without the use of oversampling, and providing digital codes tothe digital filter.

In another aspect, a digital phase locked loop circuit includes areference clock generator for providing a reference clock signal and adigital filter for providing filtered digital codes. The circuit furtherincludes a digital controlled oscillator coupled to the digital filterto receive the filtered digital codes and provide an ouput signal and afrequency divider coupled to receive the output signal and provide afrequency divided signal. Finally, the circuit includes a phase delayquantizer coupled to the frequency divider, reference clock generatorand digital filter. The phase delay quantizer is operable to quantize aphase delay between the reference clock signal and the frequency dividedsignal and provides digital codes to the digital filter based upon thesignals from the frequency divider and the reference clock generator.The phase delay quantizer quantizes the phase delay between thereference clock and the divided frequency signals without the use ofoversampling.

A digital phase delay quantizer replaces the analog charge-pump andphase frequency detector in PLL circuits. Therefore, the built-in loopfilter can be a compact-sized, high order, high bandwidth, and highattenuation digital filter as well. The digital phase locked loopcircuit takes advantage of deep sub-micron process technology whichfeatures high speed, high resolution, compact size, and low power.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates a conventional analog PLL circuit.

FIG. 2 illustrates a slower VCO adjusted by the conventional analog PLLcircuit.

FIG. 3 illustrates one embodiment of a conventional digital PLL circuit.

FIG. 4 is a block diagram of a digital PLL in accordance with anembodiment.

FIG. 5 illustrates a slower DCO adjusted by the PLL circuit of FIG. 4.

FIG. 6 illustrates the delay signal passing through each delay stage onetime only in each “f_(ref)” initiating trigger.

FIG. 7 is a diagram of an improved phase delay quantizer with a circulardelay chain.

FIG. 8 illustrates an interpretation topology which is used to achieve ashorter delay than that of a gate delay time.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS

The present invention relates to phase locked loop circuits and moreparticularly to a method for providing phase delays for such circuits.The following description is presented to enable one of ordinary skillin the art to make and use the invention and is provided in the contextof a patent application and its requirements. Various modifications tothe preferred embodiment and the generic principles and featuresdescribed herein will be readily apparent to those skilled in the art.Thus, the present invention is not intended to be limited to theembodiment shown but is to be accorded the widest scope consistent withthe principles and features described herein.

A phase locked loop circuit in accordance with an embodiment implementsa digital phase delay quantizer to replace the analog charge-pump andphase frequency detector in PLL. Therefore, the built-in loop filter canbe a compact-sized, high order, high bandwidth, and high attenuationdigital filter as well. This digital PLL circuit takes advantage of thedeep sub-micron process technology which features high speed, highresolution, compact size, and low power. To describe the features of thedigital PLL circuit in accordance with the present invention refer nowto the following description in conjunction with the accompanyingFigures.

FIG. 4 is a block diagram of a digital PLL circuit 300 in accordancewith an embodiment. Comparing the PLL circuit 300 with the conventionalanalog PLL circuit 100 shown in FIG. 1, the digital PLL circuit 300utilizes a frequency divider 312, but it replaces the phase frequencydetector and analog charge-pump with a digital phase delay quantizer304, which quantizes the phase delay between the reference clock and thedivided VCO frequency. The analog loop filter (either passive or activefilter) is replaced by a compact high order, high bandwidth, and highattenuation digital filter 308 because the input of the filter is aplurality of output digital codes from the phase delay quantizer 304.Since the output of the filter 308 are digital codes, thevoltage-controlled oscillator (VCO) of FIG. 1 is to be replaced by adigital-controlled oscillator (DCO) 310.

FIG. 5 illustrates a slower DCO 310 adjusted by the PLL circuit 300 ofFIG. 4. The phase delay quantizer 304 is employed to count the delaytime of the edges between reference clock and the divided DCO 310frequency. If the DCO 310 frequency is slower than expected, the slowerDCO 310 makes the divided DCO frequency f_(DCO)/N arrive later than thereference clock “f_(ref=)”. The delay between the two edges is bigger,and the phase delay quantizer sends the larger code(s) to the digitalfilter. The larger digital input codes for DCO 310 will increase thefrequency of the DCO 310.

To realize the function of the digital PLL circuit illustrated in FIG.5, a long delay chain based phase delay quantizer 400 is shown in FIG.6, which is a simple topology. The reference clock “f_(ref)” is sent toinitiate the delay chain, and the divided DCO frequency f_(DCO)/Narrives later to latch 412 the code that shows the propagation delaytime between the two signal edges.

However, although this system has advantages over conventional PLLcircuit, the delay chain needs to be very long to have a reasonablelocking range, which will occupy a very large silicon area and consumemore power than desired. These drawbacks can affect the ability to havecompact and low power digital circuits in deep sub-micron technology.

In FIG. 6, the delay signal 404 a-404 n passes through each delay stageone time only in each “f_(ref)” initiating trigger.

FIG. 7 is a diagram of a second embodiment delay quantizer 500 with thecircular delay chain 504 a-504 f. FIG. 8 illustrates an interpolationtopology which is used to achieve a shorter delay than that of a gatedelay time.

To describe the features of FIGS. 7 and 8 refer now to the followingdescription.

Delay ring 504 a-504 f (FIG. 7). To save the power consumption andphysical silicon size, the long delay chain was modified to form a ring504 a-504 f. Instead of counting the long delay with a very huge chain,this circular concept can dramatically reduce power consumption andcircuit size. Hence, the delay stages can be reused, the size and powerconsumption can be dramatically reduced. The digital process will countthe delay between the two edges.

Interpolating delay stages 600 (FIG. 8). To realize the fine resolution,an interpolation topology 600 (FIG. 8) is used to achieve a shorterdelay than that of a gate delay time. For example, a digital stage delayis about 15 ps in 90 nm process. If the digital delay buffer can bedivided into 4 sub-stages with interpolation structure, the delay timecan be smaller than 4 ps so as to achieve the desired resolution for ahigh performance PLL circuit. The number of interpolated sub-stages doesnot need to be 4; it can be any reasonable number (2, 3, 4, 5, 6, 7, 8 .. . ).

Code subtractor 514. The reference and divided VCO signals trigger theirindividual latch 512 respectively to record the time their edge arrives.A subtractor 514 subtracts these two codes so that it can determine thedelay time between the two edges as shown in FIG. 7. The subtracted code“R” represents the interpolated stage delays in a non-full ring circle.

Ring counter 522. If the two edge delay time is longer than that of Mbuffer stages, a ring counter 522 is used to memorize how many fullrings 520 are running between them as shown in FIG. 7. The full ringnumber “C” means that there are 4M*C interpolated stage delays.

Final code adder 516. The number of full rings 520 and the fractionalpart of the ring are summarized by a final code adder 516.

In the example shown in FIG. 7, there are M delay cells (4M interpolatedstages). For example, f_(ref) triggers latch “Latch_r” 512 a to latchthe code A(8), and after another 70 full ring delay, f_(DCO)/N triggerslatch “Latch_d” 512 b to latch the code A(15). The subtractor 514 countsthe residue R=15−8=7. Therefore, the delay time between them is known as4M*C+R=(4M*70+15−8)^(*)Δt=(208M+7)*Δt=(70 M+1+3/4)^(*)Δt_(cell), wherethe Δt_(cell) is the delay time of the delay cell and the Δt is theinterpolated stage delay time (Δt=Δt_(cell)/4 in this example).

From the above example, the benefits are obvious from the abovedescribed topologies of the phase delay quantizer 500.

(1) Due to the interpolating structure, the fine resolution is realizedwith the fractional delay time on the delay cell.

(a) Without interpolation: If M is 16 and Δt_(cell) is 16 ps, the systemknows the edge difference is between (16×70+1)×16 ps=1121×16 ps=17.936ns and (16×70+2)×16 ps=1122×16 ps=17.952 ns. Resolution is 16 ps.

(b) With interpolation: If M is 16 and Δt is 4 ps, the system knows theedge difference is between (4×16×70+7)×4 ps=4487×4 ps=17.948 ns and(4×16×70+8)×4 ps=4488×4 ps=17.952 ns. Resolution is 4 ps, which is moreprecise than the non-interpolating design.

(2) Due to the ring structure, compact size and lower power, the phasedelay quantizer 500 is realized with a circular delay ring.

(a) In this embodiment 16 delay cells are utilized instead of 1122 delaycells that would be required without the use of a ring. If a largerlocking range is desired, thousands of delay cells are required in achain structure. However, the number of delay cells are fixed when usinga circular ring structure.

Alternative Embodiments

In FIG. 5, it is assumed that the K_(DCO) is proportional to the digitalcode value (larger digital code make it faster). If the K_(DCO) isreverse proportional to the digital code value (smaller digital codemake it faster), either one of the following adjustment can keep thenegative feedback loop stable: (a) swap the two input of phase delayquantizer; (b) swap the two inputs of the subtracter in the phase delayquantizer; (c) modify the digital loop filter; and (d) modify the DCOdecoder.

The interpolating delay chain in FIG. 6 and the interpolating delay ringin FIG. 7 respectively can be either (a) single-ended design, (b)differential design, or (c) complement design.

In FIG. 7, the logic process is a conceptual graph. Some conceptualblocks can be implemented in a circuit. (a) The buffer or amplifier andlatch_r/latch_d can be integrated together (buffer or amplifier has thelatch function); (b) the latch_r/latch_d and subtracter can beintegrated together (substracter has the latch function); (c) thesubtracter and adder can be integrated together (3 input adder); (d) theadder and the result latch can be integrated together (adder has thelatch function); (e) the ring counter and the full ring stage countercan be integrated together; (f) the full ring stage counter and addercan be integrated together, and (g) the thermometer code to binary codeconverter can be neglected if the digital filter can take thethermometer code.

(h) The thermometer code to binary code converter can be placed ateither one of the digital logic process path: (i) betweenbuffer/amplifier and latch; (ii) between latch and subtracter; (iii)between subtracter and adder; (iv) between adder and result latch; (v)between result latch and digital filter and (vi) between digital filterand DCO.

Advantages

1. Compared with the conventional analog PLL circuit (FIG. 1): a highorder digital loop filter which follows the proposed digital implementedphase delay quantizer instead of the conventional analog charge-pump andthe phase frequency detector offers many advantages, such as compactsize, high-integrated level, and sharp noise attenuation. Otheradvantages are described in detail below.

(a) Fine resolution (because of interpolated delay stages). Unlike theTDC (time-to-digital converter), the proposed phase delay quantizer usesinterpolating design to achieve fine resolution so as to pursue lowphase noise, low jitter, and high linearity of the PLL performance.

(b) Large locking range with small size and lower power quantizer due tocircular concept. Unlike the TDC (time-to-digital converter), theinterpolating delay chain in the phase delay quantizer forms a ring sothat the delay stages can be reused. This circular ring dramaticallyreduces the size and power consumption of the delay chains; in addition,it offers unlimited locking range in theory.

(c) High operating frequency, and high performance with small size andlower power digital blocks (because of the low comparison rate). Unlikethe TDC (time-to-digital converter), the proposed phase delay quantizeris not an over-sampling design. Only the first stage of frequencydivider is running at the DCO output frequency, which relaxes thespeeding requirement of digital blocks; consequently, the size and powerconsumption of the digital blocks (including the proposed phase delayquantizer) are relaxed as well. The heritage of the relaxed design makesthe system generate less non-linear distortion, less jitter, and lessphase noise so as to achieve better performance. Also, this relaxeddesign has the potential to reach higher operating frequency than theconventional TDC-based digital PLL circuit.

The method and system have been described in accordance with theexemplary embodiments shown, and one of ordinary skill in the art willreadily recognize that there could be variations to the embodiments, andany variations would be within the spirit and scope of the method andsystem. Accordingly, many modifications may be made by one of ordinaryskill in the art without departing from the spirit and scope of theappended claims.

1. A method comprising: providing a reference clock signal; providingfiltered digital codes by a digital filter; receiving the filtereddigital codes and providing an output signal; receiving the outputsignal and providing a frequency divided signal; quantizing a phasedelay between the reference clock signal and the frequency dividedsignal without the use of oversampling; and providing digital codes tothe digital filter.
 2. The method of claim 1 wherein the reference clocksignal is provided by a crystal oscillator.
 3. The method of claim 1wherein the phase delay quantizer comprises a circular delay ringstructure to minimize power consumption and physical size.
 4. The methodof claim 1 wherein the quantizing step comprises providing a pluralityof interpolated delay stages to achieve a shorter delay time than a gatedelay.
 5. The method of claim 4 wherein the quantizing step includesproviding a code subtractor to subtract codes generated by the referenceclock signal and the frequency divided signal to quantize the delay timebetween edges of the signals.
 6. The method of claim 4 wherein thequantizing step includes providing a ring counter to memorize the numberof full rings that are running between buffer stages in the digitalphase locked loop circuit.